Method and apparatus for processing data for transmission in a multi-channel communication system using selective channel inversion

ABSTRACT

Techniques to process data for transmission over multiple transmission channels. The available transmission channels are segregated into one or more groups, and the channels in each group are selected for use for data transmission. Data for each group is coded and modulated based on a particular coding and modulation scheme to provide modulation symbols, and the modulation symbols for each selected channel are weighted based on an assigned weight. The weighting “inverts” the selected channels such that they achieve similar received SNRs. With selective channel inversion, only “good” channels in each group having SNRs at or above a particular threshold are selected, “bad” channels are not used, and the total available transmit power for the group is distributed across the good channels in the group. Improved performance is achieved by using only good channels in each group and matching each selected channel&#39;s received SNR to the required SNR.

CLAIM OF PRIORITY UNDER 35 U.S.C. §120

This application is a continuation of U.S. application Ser. No.09/881,610, filed Jun. 14, 2001, entitled “METHOD AND APPARATUS FORPROCESSING DATA FOR TRANSMISSION IN A MULTI CHANNEL COMMUNICATION SYSTEMUSING SELECTIVE CHANNEL INVERSION,” and assigned to the assignee hereofand hereby expressly incorporated by reference herein.

BACKGROUND

1. Field

The present invention relates generally to data communication, and morespecifically to a novel and improved method and apparatus for processingdata for transmission in a wireless communication system using selectivechannel inversion.

2. Background

A multi-channel communication system is often deployed to provideincreased transmission capacity for various types of communication suchas voice, data, and so on. Such a multi-channel system may be amultiple-input multiple-output (MIMO) communication system, anorthogonal frequency division modulation (OFDM) system, a MIMO systemthat utilizes OFDM, or some other type of system. A MIMO system employsmultiple transmit antennas and multiple receive antennas to exploitspatial diversity to support a number of spatial subchannels, each ofwhich may be used to transmit data. An OFDM system effectivelypartitions the operating frequency band into a number of frequencysubchannels (or frequency bins), each of which is associated with arespective subcarrier on which data may be modulated. A multi-channelcommunication system thus supports a number of “transmission” channels,each of which may correspond to a spatial subchannel in a MIMO system, afrequency subchannel in an OFDM system, or a spatial subchannel of afrequency subchannel in a MIMO system that utilizes OFDM.

The transmission channels of a multi-channel communication systemtypically experience different link conditions (e.g., due to differentfading and multipath effects) and may achieve differentsignal-to-noise-plus-interference ratios (SNRs). Consequently, thetransmission capacities (i.e., the information bit rates) that may besupported by the transmission channels for a particular level ofperformance may be different from channel to channel. Moreover, the linkconditions typically vary over time. As a result, the bit ratessupported by the transmission channels also vary with time.

The different transmission capacities of the transmission channels plusthe time-variant nature of these capacities make it challenging toprovide an effective coding and modulation scheme capable of processingdata prior to transmission on the channels. Moreover, for practicalconsiderations, the coding and modulation scheme should be simple toimplement and utilize at both the transmitter and receiver systems.

There is therefore a need in the art for techniques to effectively andefficiently process data for transmission on multiple transmissionchannels with different capacities.

SUMMARY

Aspects of the invention provide techniques to process data fortransmission over multiple transmission channels selected from among allavailable transmission channels. The available transmission channels(e.g., the spatial subchannels and frequency subchannels in a MIMOsystem that utilizes OFDM) are segregated into one or more groups, witheach group including any number of transmission channels. In an aspect,the data processing includes coding and modulating data for each groupbased on a common coding and modulation scheme selected for that groupto provide modulation symbols and weighting the modulation symbols foreach selected transmission channel based on a weight assigned to thechannel. The weighting effectively “inverts” the selected transmissionchannels in each group such that these channels achieve approximatelysimilar received signal-to-noise-plus-interference ratios (SNRs).

In one embodiment, which is referred to as selective channel inversion(SCI), only “good” transmission channels in each group having SNRs (orpower gains) at or above a particular (SNR or power gain) threshold areselected for use for data transmission, and “bad” transmission channelsare not used. With selective channel inversion, the total availabletransmit power for each group is distributed (unevenly) across the goodtransmission channels, and improved efficiency and performance areachieved. In another embodiment, all available transmission channels ineach group are selected for use and the channel inversion is performedfor all available channels in the group.

Each group of transmission channels may be associated with (1) arespective (SNR or power gain) threshold used to select transmissionchannels for use for data transmission and (2) a respective coding andmodulation scheme used to code and modulate the data for the group. Fora MIMO system that utilizes OFDM, each group may correspond to arespective transmit antenna, and the transmission channels in each groupmay be the frequency subchannels for the corresponding transmit antenna.

The channel inversion techniques simplify the coding/modulation at atransmitter system and the decoding/demodulation at a receiver system.Moreover, the selective channel inversion technique may also provideimproved performance due to the combined benefits of (1) using only theN_(S) best transmission channels in each group selected from among allavailable transmission channels in the group and (2) matching thereceived SNR of each selected transmission channel to the SNR requiredby the coding and modulation scheme used for the group in which thechannel belongs.

The invention further provides methods, systems, and apparatus thatimplement various aspects, embodiments, and features of the invention,as described in further detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, nature, and advantages of the present invention willbecome more apparent from the detailed description set forth below whentaken in conjunction with the drawings in which like referencecharacters identify correspondingly throughout and wherein:

FIG. 1 is a diagram of a multiple-input multiple-output (MIMO)communication system that may be designed and operated to implementvarious aspects and embodiments of the invention;

FIG. 2A is a flow diagram of a process to determine the amount oftransmit power to be allocated to each selected transmission channelbased on selective channel inversion, in accordance with an embodimentof the invention;

FIG. 2B is a flow diagram of a process to determine a threshold a usedto select transmission channels for data transmission, in accordancewith an embodiment of the invention;

FIG. 3 is a diagram of a MIMO communication system capable ofimplementing various aspects and embodiments of the invention;

FIGS. 4A through 4D are block diagrams of four MIMO transmitter systemscapable of processing data in accordance with four specific embodimentsof the invention;

FIG. 5 is a block diagrams of a MIMO receiver system capable ofreceiving data in accordance with an embodiment of the invention;

FIGS. 6A and 6B are block diagrams of an embodiment of a channelMIMO/data processor and an interference canceller, respectively, withinthe MIMO receiver system shown in FIG. 5; and

FIG. 7 is a block diagram of a MIMO receiver system capable of receivingdata in accordance with another embodiment of the invention.

DETAILED DESCRIPTION

Various aspects, embodiments, and features of the invention may beapplied to any multi-channel communication system in which multipletransmission channels are available for data transmission. Suchmulti-channel communication systems include multiple-inputmultiple-output (MIMO) systems, orthogonal frequency division modulation(OFDM) systems, MIMO systems that utilize OFDM, and others. Themulti-channel communication systems may also implement code divisionmultiple access (CDMA), time division multiple access (TDMA), frequencydivision multiple access (FDMA), or some other multiple accesstechniques. Multiple access communication systems can support concurrentcommunication with a number of terminals (i.e., users).

FIG. 1 is a diagram of a multiple-input multiple-output (MIMO)communication system 100 that may be designed and operated to implementvarious aspects and embodiments of the invention. MIMO system 100employs multiple (N_(T)) transmit antennas and multiple (N_(R)) receiveantennas for data transmission. MIMO system 100 is effectively formedfor a multiple access communication system having a base station (BS)104 that concurrently communicates with a number of terminals (T) 106.In this case, base station 104 employs multiple antennas and representsthe multiple-input (MI) for uplink transmissions and the multiple-output(MO) for downlink transmissions. The downlink (i.e., forward link)refers to transmissions from the base station to the terminals, and theuplink (i.e., reverse link) refers to transmissions from the terminalsto the base station.

A MIMO system employs multiple (N_(T)) transmit antennas and multiple(N_(R)) receive antennas for data transmission. A MIMO channel formed bythe N_(T) transmit and N_(R) receive antennas may be decomposed intoN_(C) independent channels, with N_(C)≦min {N_(T), N_(R)}. Each of theN_(C) independent channels is also referred to as a spatial subchannelof the MIMO channel and corresponds to a dimension. In one common MIMOsystem implementation, the N_(T) transmit antennas are located at andassociated with a single transmitter system, and the N_(R) receiveantennas are similarly located at and associated with a single receiversystem. A MIMO system may also be effectively formed for a multipleaccess communication system having a base station that concurrentlycommunicates with a number of terminals. In this case, the base stationis equipped with a number of antennas and each terminal may be equippedwith one or more antennas.

An OFDM system effectively partitions the operating frequency band intoa number of (N_(F)) frequency subchannels (i.e., frequency bins orsubbands). At each time slot, a modulation symbol may be transmitted oneach of the N_(F) frequency subchannels. Each time slot corresponds to aparticular time interval that may be dependent on the bandwidth of thefrequency subchannel.

A multi-channel communication system may be operated to transmit datavia a number of transmission channels. For a MIMO system not utilizingOFDM, there is typically only one frequency subchannel and each spatialsubchannel may be referred to as a transmission channel. For a MIMOsystem utilizing OFDM, each spatial subchannel of each frequencysubchannel may be referred to as a transmission channel. And for an OFDMsystem not utilizing MIMO, there is only one spatial subchannel for eachfrequency subchannel and each frequency subchannel may be referred to asa transmission channel.

The transmission channels in a multi-channel communication systemtypically experience different link conditions (e.g., due to differentfading and multipath effects) and may achieve differentsignal-to-noise-plus-interference ratios (SNRs). Consequently, thecapacity of the transmission channels may be different from channel tochannel. This capacity may be quantified by the information bit rate(i.e., the number of information bits per modulation symbol) that may betransmitted on a transmission channel for a particular level ofperformance (e.g., a particular bit error rate (BER) or packet errorrate (PER)). Since the link conditions typically vary with time, thesupported information bit rates for the transmission channels also varywith time.

To more fully utilize the capacity of the transmission channels, channelstate information (CSI) descriptive of the link conditions may bedetermined (typically at the receiver system) and provided to thetransmitter system. The transmitter system may then process (e.g.,encode, modulate, and weight) data such that the transmitted informationbit rate for each transmission channel matches the transmission capacityof the channel. CSI may be categorized as either “full CSI” or “partialCSI”. Full CSI includes sufficient characterization (e.g., the amplitudeand phase) across the entire system bandwidth for the propagation pathbetween each transmit-receive antenna pair in a N_(T)×N_(R) MIMO matrix(i.e., the characterization for each transmission channel). Partial CSImay include, for example, the SNRs of the transmission channels.

Various techniques may be used to process data prior to transmissionover multiple transmission channels. In one technique, data for eachtransmission channel may be coded and modulated based on a particularcoding and modulation scheme selected for that channel based on thechannel's CSI. By coding and modulating separately for each transmissionchannel, the coding and modulation may be optimized for the SNR achievedby each channel. In one implementation of such a technique, a fixed basecode is used to encode data, and the coded bits for each transmissionchannel are then punctured (i.e., selectively deleted) to obtain a coderate supported by that channel. In this implementation, the modulationscheme for each transmission channel is also selected based on thechannel's code rate and SNR. This coding and modulation scheme isdescribed in further detail in U.S. patent application Ser. No.09/776,075, entitled “CODING SCHEME FOR A WIRELESS COMMUNICATIONSYSTEM,” filed Feb. 1, 2001, assigned to the assignee of the presentapplication and incorporated herein by reference. For this technique,substantial implementation complexity is typically associated withhaving a different code rate and modulation scheme for each transmissionchannel.

In accordance with an aspect of the invention, techniques are providedto (1) process data for all selected transmission channels based on acommon coding and modulation scheme to provide modulation symbols, and(2) weight the modulation symbols for each selected transmission channelbased on the channel's CSI. The weighting effectively “inverts” theselected transmission channels such that, in general, the SNRs areapproximately similar at the receiver system for all selectedtransmission channels. In one embodiment, which is referred to asselective channel inversion (SCI), only “good” transmission channelshaving SNRs (or power gains) at or above a particular SNR (or powergain) threshold are selected for use for data transmission, and “bad”transmission channels are not used. With selective channel inversion,the total available transmit power is distributed across the goodtransmission channels, and improved efficiency and performance areachieved. In another embodiment, all available transmission channels areselected for use and the channel inversion is performed for alltransmission channels.

In yet another embodiment, the available transmission channels aresegregated into groups and the selective channel inversion is appliedindependently to each group of channels. For example, the frequencysubchannels of each transmit antenna may be grouped together, and theselective channel inversion may be applied independently for each of thetransmit antennas. This segregation permits the optimization to beachieved on a per group (e.g., per transmit antenna) basis.

These channel inversion techniques may be advantageously used when fullor partial CSI is available at the transmitter. These techniquesameliorate most of the complexity associated with the channel-specificcoding and modulation technique described above, while still achievinghigh performance. Moreover, the selective channel inversion techniquemay also provide improved performance over the channel-specific codingand modulation technique due to the combined benefits of (1) using onlythe N_(S) best transmission channels from among the availabletransmission channels and (2) matching the received SNR of each selectedtransmission channel to the SNR required for the selected coding andmodulation scheme.

For a MIMO system utilizing OFDM and having full CSI available, thetransmitter system may have knowledge of the complex-valued gain of thetransmission path between each transmit-receive antenna pair of eachfrequency subchannel. This information may be used to render the MIMOchannel orthogonal so that each eigenmode (i.e., spatial subchannel) maybe used for an independent data stream.

For a MIMO system utilizing OFDM and having partial CSI available, thetransmitter may have limited knowledge of the transmission channels.Independent data streams may be transmitted on correspondingtransmission channels over the available transmit antennas, and thereceiver system may use a particular linear (spatial) or non-linear(space-time) processing technique (i.e., equalization) to separate outthe data streams. The equalization provides an independent data streamcorresponding to each transmission channel (e.g., each transmit antennaand/or each frequency subchannel), and each of these data streams has anassociated SNR.

If the set of SNRs for the transmission channels is available at thetransmitter system, this information may be used to select the propercoding and modulation scheme and to distribute the total availabletransmit power for each group (there may be only one group). In anembodiment, the available transmission channels in each group are rankedin order of decreasing received SNR, and the total available transmitpower is allocated to and used for the N_(S) best transmission channelsin the group. In an embodiment, transmission channels having receivedSNRs that fall below a particular SNR threshold are not selected foruse. The SNR threshold may be selected to optimize throughput or someother criteria. The total available transmit power for each group isdistributed across all transmission channels in the group selected foruse such that the transmitted data streams have approximately similarreceived SNRs at the receiver system. Similar processing may beperformed if the channel gains are available at the transmitter system.In an embodiment, a common coding scheme (e.g., a particular Turbo codeof a particular code rate) and a common modulation scheme (e.g., aparticular PSK or QAM constellation) are used for all selectedtransmission channels in each group.

Transmission Channel Inversion

If a simple (common) coding and modulation scheme can be used at thetransmitter system, then a single (e.g., convolutional or Turbo) coderand code rate may be used to encode data for all transmission channelsselected for data transmission, and the resultant coded bits may bemapped to modulation symbols using a single (e.g., PSK or QAM)modulation scheme. The resultant modulation symbols are then all drawnfrom the same “alphabet” of possible modulation symbols and encoded withthe same code and code rate. This would then simplify the dataprocessing at both the transmitter and receiver.

However, the transmission channels in a multi-channel communicationsystem typically experience different link conditions and achievedifferent SNRs. In this case, if the same amount of transmit power isused for each selected transmission channel, then the transmittedmodulation symbols will be received at different SNRs depending on thespecific channels on which the modulation symbols are transmitted. Theresult may be a large variation in symbol error probability over the setof selected transmission channels, and an associated loss in bandwidthefficiency.

In accordance with an aspect of the invention, a power control mechanismis used to set or adjust the transmit power level for each transmissionchannel selected for data transmission to achieve a particular SNR atthe receiver system. By achieving similar received SNRs for all selectedtransmission channels, a single coding and modulation scheme may be usedfor all selected transmission channels, which can greatly reduce thecomplexity of the coding/modulation process at the transmitter systemand the complementary demodulation/decoding process at the receiversystem. The power control may be achieved by “inverting” the selectedtransmission channels and properly distributing the total availabletransmit power across all selected channels, as described in furtherdetail below.

If the same amount of transmit power is used for all availabletransmission channels in a MIMO system utilizing OFDM, then the receivedpower for a particular channel may be expressed as: $\begin{matrix}{{P_{rx}^{\prime}\left( {j,k} \right)} = {\frac{P_{tx}}{N_{T}N_{F}}{{H\left( {j,k} \right)}}^{2}}} & {{Eq}\quad(1)}\end{matrix}$where

-   -   P_(rx)′(j,k) is the received power for transmission channel        (j,k) (i.e., the j-th spatial subchannel of the k-th frequency        subchannel),    -   P_(tx) is the total transmit power available at the transmitter,    -   N_(T) is the number of transmit antennas,    -   N_(F) is the number of frequency subchannels, and    -   H(j,k) is the complex-valued “effective” channel gain from the        transmitter to the receiver for transmission channel (j,k).        For simplicity, the channel gain H(j,k) includes the effects of        the processing at the transmitter and receiver. Also for        simplicity, it is assumed that the number of spatial subchannels        is equal to the number of transmit antennas and N_(T)·N_(F)        represents the total number of available transmission channels.        If the same amount of power is transmitted for each available        transmission channel, the total received power P_(rx) _(—)        _(total) for all available transmission channels may be        expressed as: $\begin{matrix}        {P_{rx\_ total} = {\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{\frac{P_{tx}}{N_{T}N_{F}}{{{H\left( {j,k} \right)}}^{2}.}}}}} & {{Eq}\quad(2)}        \end{matrix}$

Equation (1) shows that the receive power for each transmission channelis dependent on the power gain of that channel, i.e., |H(j,k)|². Toachieve equal received power across all available transmission channels,the modulation symbols for each channel can be weighted at thetransmitter by a weight of W(j,k), which can be expressed as:$\begin{matrix}{{{W\left( {j,k} \right)} = \frac{c}{{H\left( {j,k} \right)}}},} & {{Eq}\quad(3)}\end{matrix}$where c is a factor chosen such that the received powers for alltransmission channels are approximately equal at the receiver. As shownin equation (3), the weight for each transmission channel is inverselyproportional to that channel's gain. The weighted transmit power fortransmission channel (j,k) can then be expressed as: $\begin{matrix}{{{P_{tx}\left( {j,k} \right)} = \frac{{bP}_{tx}}{{{H\left( {j,k} \right)}}^{2}}},} & {{Eq}\quad(4)}\end{matrix}$where b is a “normalization” factor used to distribute the totaltransmit power among the available transmission channels. Thisnormalization factor b can be expressed as: $\begin{matrix}{{b = \frac{1}{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{H\left( {j,k} \right)}}^{- 2}}}},} & {{Eq}\quad(5)}\end{matrix}$where c²=b. As shown in equation (5), the normalization factor b iscomputed as the sum of the reciprocal power gains for all availabletransmission channels.

The weighting of the modulation symbols for each transmission channel byW(j,k) effectively “inverts” the transmission channel. This channelinversion results in the amount of transmit power for each transmissionchannel being inversely proportional to the channel's power gain, asshown in equation (4), which then provides a particular received powerat the receiver. The total available transmit power is thus effectivelydistributed (unevenly) to all available transmission channels based ontheir channel gains such that all transmission channels haveapproximately equal received power, which may be expressed as:P _(rx)(j,k)=bP _(tx).  Eq (6)If the noise variance is the same across all transmission channels, thenthe equal received power allows the modulation symbols for all channelsto be generated based on a single common coding and modulation scheme,which then greatly simplify the coding and decoding processes.

If all available transmission channels are used for data transmissionregardless of their channel gains, then the poor transmission channelsare allocated more of the total transmit power. In fact, to achievesimilar received power for all transmission channels, the poorer atransmission channel gets the more transmit power needs to be allocatedto this channel. When one or more transmission channels becomeexcessively poor, the amount of transmit power needed for these channelswould deprive (or starve) the good channels of power, which may thendramatically decrease the overall system throughput.

Selective Channel Inversion Based on Channel Gains

In an aspect, the channel inversion is applied selectively, and onlytransmission channels whose received power is at or above a particularthreshold, α, relative to the total received power are selected for datatransmission. Transmission channels whose received power falls belowthis threshold are erased (i.e., not used). For each selectedtransmission channel, the modulation symbols are weighted at thetransmitter such that all selected transmission channels are received atapproximately similar power level. The threshold can be selected tomaximize throughput or based on some other criteria. The selectivechannel inversion scheme preserves most of the simplicity inherent inusing a common coding and modulation scheme for all transmissionchannels while also provides high performance normally associated withindividual coding per transmission channel.

Initially, the average power gain, L_(ave), is computed for allavailable transmission channels and can be expressed as: $\begin{matrix}{L_{ave} = {\frac{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{H\left( {j,k} \right)}}^{2}}}{N_{T}N_{F}}.}} & {{Eq}\quad(7)}\end{matrix}$

The modulation symbols for each selected transmission channel can beweighted at the transmitter by a weight of {tilde over (W)}(j,k), whichcan be expressed as: $\begin{matrix}{{\overset{\sim}{W}\left( {j,k} \right)} = {\frac{\overset{\sim}{c}}{{H\left( {j,k} \right)}}.}} & {{Eq}\quad(8)}\end{matrix}$The weight for each selected transmission channel is inverselyproportional to that channel's gain and is determined such that allselected transmission channels are received at approximately equalpower. The weighted transmit power for each transmission channel canthen be expressed as: $\begin{matrix}{{P_{tx}\left( {j,k} \right)} = \left\{ {\begin{matrix}{\frac{\overset{\sim}{b}P_{tx}}{{{H\left( {j,k} \right)}}^{2}},} & {{{H\left( {j,k} \right)}}^{2} \geq {\alpha\quad L_{ave}}} \\{0,} & {otherwise}\end{matrix},} \right.} & {{Eq}\quad(9)}\end{matrix}$where α is the threshold and {tilde over (b)} is a normalization factorused to distribute the total transmit power among the selectedtransmission channels. As shown in equation (9), a transmission channelis selected for use if its power gain is greater than or equal to apower gain threshold (i.e., |H(j,k)|²≧αL_(ave)) The normalization factor{tilde over (b)} is computed based on only the selected transmissionchannels and can be expressed as: $\begin{matrix}{\overset{\sim}{b} = {\frac{1}{\sum\limits_{{{H{({j,k})}}}^{2} \geq {\alpha\quad L_{ave}}}{{H\left( {j,k} \right)}}^{- 2}}.}} & {{Eq}\quad(10)}\end{matrix}$

Equations (7) through (10) effectively distribute the total transmitpower to the selected transmission channels based on their power gainssuch that all selected transmission channels have approximately equalreceived power, which may be expressed as: $\begin{matrix}{{P_{rx}\left( {j,k} \right)} = \left\{ {\begin{matrix}{{\overset{\sim}{b}P_{tx}},} & {{{H\left( {j,k} \right)}}^{2} \geq {\alpha\quad L_{ave}}} \\{0,} & {otherwise}\end{matrix}.} \right.} & {{Eq}\quad(11)}\end{matrix}$

Selective Channel Inversion Based on Channel SNRs

In many communication systems, the known quantities at the receiversystem are the received SNRs for the transmission channels rather thanthe channel gains (i.e., the path losses). In such systems, theselective channel inversion technique can be readily modified to operatebased on the received SNRs instead of the channel gains.

If equal transmit power is used for all available transmission channelsand the noise variance, σ², is constant for all channels, then thereceived SNR, γ(j,k), for transmission channel (j,k) can be expressedas: $\begin{matrix}{{\gamma\left( {j,k} \right)} = {\frac{P_{{rx}\quad}\left( {j,k} \right)}{\sigma^{2}} = {\frac{P_{tx}}{\sigma^{2}N_{T\quad}N_{F}}{{{H\left( {j,k} \right)}}^{2}.}}}} & {{Eq}\quad(12)}\end{matrix}$The average received SNR, γ_(ave), for each available transmissionchannel may be expressed as: $\begin{matrix}{{\gamma_{ave} = {\frac{P_{tx}}{{\sigma^{2}\left( {N_{T\quad}N_{F}} \right)}^{2}}{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{H\left( {j,k} \right)}}^{2}}}}},} & {{Eq}\quad(13)}\end{matrix}$which also assumes equal transmit power over the available transmissionchannels. The received SNR, γ_(total), for all available transmissionchannels may be expressed as: $\begin{matrix}{\gamma_{total} = {{\frac{P_{tx}}{\sigma^{2}}L_{ave}} = {\frac{P_{tx}}{\sigma^{2}N_{T\quad}N_{F}}{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{{H\left( {j,k} \right)}}^{2}.}}}}}} & {{Eq}\quad(14)}\end{matrix}$The total received SNR, γ_(total), is based on the total transmit powerbeing equally distributed across all available transmission channels.

A normalization factor, β, used to distribute the total transmit poweramong the selected transmission channels can be expressed as:$\begin{matrix}{\beta = {\frac{1}{\sum\limits_{{\gamma{({j,k})}} \geq {\alpha\quad\gamma_{ave}}}^{\quad}{\gamma\left( {j,k} \right)}^{- 1}}.}} & {{Eq}\quad(15)}\end{matrix}$As shown in equation (15), the normalization factor β is computed basedon, and as the sum of the reciprocal of, the SNRs of all selectedtransmission channels.

To achieve similar received SNR for all selected transmission channels,the modulation symbols for each selected transmission channel (j,k) maybe weighted by a weight that is related to that channel's SNR, which maybe expressed as: $\begin{matrix}{{\overset{\sim}{W}\left( {j,k} \right)} = {\frac{\overset{\sim}{c}}{\sqrt{\gamma\left( {j,k} \right)}}.}} & {{Eq}\quad(16)}\end{matrix}$where {tilde over (c)}²=β. The weighted transmit power for eachtransmission channel may then be expressed as: $\begin{matrix}{{P_{tx}\left( {j,k} \right)} = \left\{ {\begin{matrix}{\frac{\beta\quad P_{tx}}{\gamma\left( {j,k} \right)},} & {{\gamma\left( {j,k} \right)} \geq {\alpha\quad\gamma_{ave}}} \\{0,} & {otherwise}\end{matrix}.} \right.} & {{Eq}\quad(17)}\end{matrix}$As shown in equation (17), only transmission channels for which thereceived SNR is greater than or equal to an SNR threshold (i.e.,γ(j,k)≧αγ_(ave)) are selected for use.

If the total transmit power is distributed across all selectedtransmission channels such that the received SNR is approximatelysimilar for all selected channels, then the resulting received SNR foreach transmission channel may be expressed as: $\begin{matrix}{{\overset{\sim}{\gamma}\left( {j,k} \right)} = \left\{ {\begin{matrix}{\frac{\beta\quad\gamma_{total}}{\gamma_{ave}},} & {{\gamma\left( {j,k} \right)} \geq {\alpha\quad\gamma_{ave}}} \\{0,} & {otherwise}\end{matrix}.} \right.} & {{Eq}\quad(18)}\end{matrix}$By substituting γ_(ave) from equation (13) and γ_(total) from equation(14) into equation (18), the following is obtained:${\overset{\sim}{\gamma}\left( {j,k} \right)} = \left\{ {\begin{matrix}{{\beta\quad N_{T}N_{F}},} & {{\gamma\left( {j,k} \right)} \geq {\alpha\quad\gamma_{ave}}} \\{0,} & {otherwise}\end{matrix}.} \right.$

Channel Inversion for Segregated Groups of Transmission Channels

In the above description, the channel inversion is applied to allavailable transmission channels or selectively to a subset of theavailable transmission channels (which are selected based on aparticular threshold). This then allows a common coding and modulationscheme to be used for all transmission channels to be used for datatransmission.

The selective channel inversion may also be applied individually andindependently to groups of transmission channels. In this case, theavailable transmission channels in the communication system areinitially segregated into a number of groups. Any number of groups maybe formed, and each group may include any number of channels (i.e.,there need not be equal number of channels in each group).

A particular amount of transmit power is also available for each groupbased on various system constraints and considerations. For a fullchannel inversion technique, the available transmit power for each groupis allocated to all transmission channels in the group such that thereceived signal quality for these channels is approximately equal (i.e.,similar received SNRs). And for a selective channel inversion technique,all or a subset of the available transmission channels in each group areselected for use, e.g., based on a particular threshold determined forthe group. The available transmit power for each group is then allocatedto the selected transmission channels in the group such that thereceived signal quality for the channels is approximately equal.

Various additional flexibilities are afforded by processing dataseparately for each group of transmission channels. For example, thefull or selective channel inversion may be independently applied to eachgroup of channels. Also, for those groups for which selective channelinversion is applied, one threshold may be used for all groups, eachgroup may be assigned a separate threshold, or some groups may share thesame threshold while other groups may be assigned separate thresholds. Adifferent coding and modulation scheme may also be used for each group,which may be selected based on the received SNR achieved by thetransmission channels in the group.

For a MIMO system that utilizes OFDM, the MIMO construct createsmultiple (N_(S)) transmission channels in the spatial domain and theOFDM construct creates multiple (N_(F)) transmission channels in thefrequency domain. The total number of transmission channels available tosend data is then N=N_(S)·N_(F). The N transmission channels may then besegregated into a number of groups in various ways.

In one embodiment, the transmission channels are segregated on a pertransmit antenna basis. If the number of spatial subchannels is equal tothe number of transmit antennas (i.e., N_(T)=N_(S)), then the full orselective channel inversion may be applied independently to each of theN_(T) transmit antennas. In an embodiment, selective channel inversionis used for each group, and the N_(T) groups corresponding to the N_(T)transmit antennas may be associated with N_(T) respective thresholds,one threshold for each group or transmit antenna. The selective channelinversion then determines the subset of transmission channels (orfrequency subchannels) associated with each transmit antenna havingadequate received SNRs, which can be achieved by comparing the receivedSNR for each frequency subchannel to the threshold for the transmitantenna. The total transmit power available for each transmit antenna isthen allocated to the selected frequency subchannels for the transmitantenna such that the received SNRs for these frequency subchannels areapproximately similar.

In another embodiment, the available transmission channels aresegregated on a per frequency subchannel basis. In this embodiment, thefull or selective channel inversion may be applied independently to eachof the N_(F) frequency subchannels. If selective channel inversion isused, then the spatial subchannels in each group may be selected for usefor data transmission based on the threshold for the group correspondingto that frequency subchannel.

The segregation of the available transmission channels into groupspermits optimization to be achieved on a per group basis (e.g., pertransmit antenna or per frequency subchannel), which then allows aspecific coding and modulation scheme to be used for all selectedtransmission channels in each group. For example, one or more transmitantennas may be assigned to each scheduled terminal for datatransmission. The transmission channels associated with the assignedtransmit antennas may be placed in a group, and the selective channelinversion may be performed on this group of transmission channels suchthat a single coding and modulation scheme may be used for the datatransmission to this terminal.

If equal transmit power is used for all available transmission channelsin group j and the noise variance, σ², is constant for all channels,then the received SNR, γ_(j)(k), for transmission channel k in group jcan be expressed as: $\begin{matrix}{{\gamma_{j}(k)} = {\frac{P_{{rx},j}(k)}{\sigma^{2}} = {\frac{P_{{tx},j}}{\sigma^{2}N_{j}}{{{H_{j}(k)}}^{2}.}}}} & {{Eq}\quad(19)}\end{matrix}$where

-   -   P_(rx,j)(k) is the received power for transmission channel k in        group j,    -   P_(tx,j) is the total available transmit power for group j,    -   H_(j)(k) is effective channel gain from the transmitter to the        receiver for transmission channel k in group j, and    -   N_(j) is the number of transmission channels in group j. Group j        may correspond to a specific transmit antenna j, in which case        N_(j)=N_(F).        The average received SNR, γ_(ave,j), for each available        transmission channel in group j may be expressed as:        $\begin{matrix}        {\gamma_{{ave},j} = {\frac{P_{{tx},j}}{\sigma^{2}N_{j}^{2}}{\sum\limits_{k = 1}^{N_{j}}{{{H_{j}(k)}}^{2}.}}}} & {{Eq}\quad(20)}        \end{matrix}$        Equation (20) assumes equal transmit power over the N_(j)        available transmission channels in group j. The received SNR,        γ_(total,j), for all available transmission channels in group j        may then be expressed as: $\begin{matrix}        {{\gamma_{{total},j} = {{\frac{P_{{tx},j}}{\sigma^{2}}L_{{ave},j}} = {\frac{P_{{tx},j}}{\sigma^{2}N_{j}}{\sum\limits_{k = 1}^{N_{j}}{{H_{j}(k)}}^{2}}}}},} & {{Eq}\quad(21)}        \end{matrix}$        where $\begin{matrix}        {L_{{ave},j} = {\frac{1}{N_{j}}{\sum\limits_{k = 1}^{N_{j}}{{{H_{j}(k)}}^{2}.}}}} & {{Eq}\quad(22)}        \end{matrix}$        The total received SNR, γ_(total,j), for group j is based on the        total transmit power, P_(tx,j), for group j being equally        distributed across all available transmission channels in the        group.

A normalization factor, β_(j), used to distribute the total transmitpower P_(tx,j) among the selected transmission channels in group j canbe expressed as: $\begin{matrix}{\beta_{j} = {\frac{1}{\sum\limits_{{\lambda_{j}{(k)}} \geq {\alpha_{j}\lambda_{{ave},j}}}{\gamma_{j}(k)}^{- 1}}.}} & {{Eq}\quad(23)}\end{matrix}$As shown in equation (23), the normalization factor β_(j) is computedbased on the SNRs of all selected transmission channels in group j, withthe channels being selected based on the threshold, α_(j)γ_(ave,j),determined for the group.

To achieve similar received SNR for all selected transmission channelsin the group, the modulation symbols for each selected transmissionchannel may be weighted by a weight that is related to that channel'sSNR, which may be expressed as: $\begin{matrix}{{{\overset{\sim}{W}}_{j}(k)} = {\frac{\overset{\sim}{c}}{\sqrt{\gamma_{j}(k)}}.}} & {{Eq}\quad(24)}\end{matrix}$where {tilde over (c)}_(j) ²=β_(j). The weighted transmit power for eachtransmission channel may then be expressed as: $\begin{matrix}{{P_{{tx},j}(k)} = \left\{ {\begin{matrix}\frac{\beta_{j}P_{{tx},j}}{\gamma_{j}(k)} & {,{{\gamma_{j}(k)} \geq {\alpha_{j}\gamma_{{ave},j}}}} \\0 & {,{otherwise}}\end{matrix}.} \right.} & {{Eq}\quad(25)}\end{matrix}$As shown in equation (25), only transmission channels for which thereceived SNR is greater than or equal to the SNR threshold (i.e.,γ_(j)(k)≧α_(j)γ_(ave,j)) are selected for use.

If the total transmit power is distributed across all selectedtransmission channels in the group such that the received SNR isapproximately similar for all selected channels, then the resultingreceived SNR for each transmission channel may be expressed as:$\begin{matrix}{{{\overset{\sim}{b}(\ell)} = \frac{1}{\sum\limits_{i = 1}^{\ell}{{H\left( {j,k} \right)}}^{- 2}}},{1 \leq \ell \leq {N_{T}{N_{F}.}}}} & {{Eq}\quad(27)}\end{matrix}$

The process described above may be repeated for each group oftransmission channels. Each group may be associated with a differentthreshold, α_(j)γ_(ave,j), derived to provide the desire performance forthat group. The ability to allocate transmit power on a per group (e.g.,per transmit antenna) basis can provide enhanced flexibility and mayfurther improve performance.

FIG. 2A is a flow diagram of a process 200 to determine the amount oftransmit power to be allocated to each selected transmission channelbased on selective channel inversion, in accordance with an embodimentof the invention. Process 200 assumes that all available transmissionchannels are considered (i.e., one group of transmission channels forthe communication system). Process 200 may be used if the channel gainsH(j,k), the received SNRs γ(j,k), or some other characteristics areavailable for the transmission channels. For clarity, process 200 isdescribed below for the case in which the channel gains are available,and the case in which the received SNRs are available is shown withinbrackets.

Initially, the channel gains H(j,k) [or the received SNRs γ(j,k)] of allavailable transmission channels are retrieved, at step 212. A power gainthreshold, αL_(ave), [or an SNR threshold, αγ_(ave)] used to selecttransmission channels for data transmission is also determined, at step214. The threshold may be computed as described in further detail below.

Each available transmission channel is then evaluated for possible use.A (not yet evaluated) available transmission channel is identified forevaluation, at step 216. For the identified transmission channel, adetermination is made whether or not the power gain [or the receivedSNR] for the channel is greater than or equal to the power gainthreshold (i.e., |H(j,k)|²≧αL_(ave)) [or the SNR threshold (i.e.,γ(j,k)≧αγ_(ave)], at step 218. If the identified transmission channelsatisfies the criteria, then it is selected for use, at step 220.Otherwise, if the transmission channel does not satisfy the criteria, itis discarded and not used for data transmission.

A determination is then made whether or not all available transmissionchannels have been evaluated, at step 222. If not, the process returnsto step 216 and another available transmission channel is identified forevaluation. Otherwise, the process proceeds to step 224.

At step 224, a normalization factor {tilde over (b)} [or β] used todistribute the total transmit power among the selected transmissionchannels is determined based on the channel gains [or the received SNRs]of the selected channels, at step 224. This can be achieved as shown inequation (10) [or equation (15)]. A weight {tilde over (W)}(j,k) is nextcomputed for each selected transmission channel, at step 226, based onthe normalization factor and that channel's gain [or SNR]. The weightcan be computed as shown in equation (8) [or equation (16)]. Theweighted transmit power for each selected transmission channel wouldthen be as shown in equation (9) [or equation (17)]. The process thenterminates.

In the above description, the total available transmit power for eachgroup is allocated (unevenly) to the selected transmission channels inthe group based on their respective weights such that the received SNRsfor these channels are approximately similar. (There may be only onegroup of transmission channels.) In some other embodiments, the totalavailable transmit power may be allocated equally amongst the selectedtransmission channels, in which case the weights for the selectedtransmission channels are equal. This may be implemented, for example,if the common coding and modulation scheme for a group is selected basedon the average SNR for the selected transmission channels in the group.The desired level of performance may be achieved, for example, byinterleaving the data across all selected transmission channels in thegroup or via some other processing scheme.

Threshold Selection

The threshold, α, used to select transmission channels for use for datatransmission may be set based on various criteria. In one embodiment,the threshold is set to optimize throughput.

Initially, a vector of setpoints (i.e., Z=[z₁, z₂, . . . , z_(N) _(Z) ])and a vector of code rates (i.e., R=[r₁, r₂, . . . , r_(N) _(Z) ]) aredefined. The code rates include the effects of the coding and modulationscheme and are representative of the number of information bits permodulation symbol. Each vector includes N_(Z) elements corresponding tothe number of available code rates, which may be those available for usein the system. Alternatively, N_(Z) setpoints may be defined based onthe operating points supported by the system. Each setpoint correspondsto a particular received SNR needed to achieve a particular level ofperformance. The setpoint is typically dependent on the transmission bitrate (i.e., the number of information bits per modulation symbol), whichis further dependent on the code rate and the modulation scheme used forthe data transmission. As noted above, a common modulation scheme isused for all selected transmission channels. In this case, thetransmission bit rate and thus the setpoint is directly related to thecode rate.

Each code rate r_(n), where 1≦n≦N_(Z), is associated with a respectivesetpoint z_(n), which is the minimum received SNR required to operate atthat code rate for the required level of performance. The requiredsetpoint z_(n) may be determined based on computer simulation,mathematical derivation, and/or empirical measurement, as is known inthe art. The elements in the two vectors R and Z may also be orderedsuch that {z₁>z₂> . . . >z_(N) _(Z) } and {r₁>r₂> . . . >r_(N) _(Z) },with z₁ being the largest setpoint and r₁ being the highest supportedcode rate.

The channel gains for all available transmission channels are used tocompute power gains, which are then ranked and placed in a list H(l) inorder of decreasing power gains, where 1≦l≦N_(T)N_(F), such thatH(1)=max {|H(j,k)|²}, . . . , and H(N_(T)N_(F))=min {|H(j,k)|²}.

A sequence {tilde over (b)}(l) of possible normalization factors is alsodefined as follows: $\begin{matrix}{{\frac{{\overset{\sim}{b}(\ell)}P_{tx}}{\sigma^{2}} \geq z_{n}},} & {{Eq}\quad(28)}\end{matrix}$Each element of the sequence {tilde over (b)}(l) may be used as anormalization factor if the l best transmission channels are selectedfor use.

For each code rate r_(n) (where 1≦n≦N_(Z)), the largest value of l,l_(n,max), is determined such that the received SNR for each of the lbest transmission channels is greater than or equal to the setpointz_(n) associated with the code rate r_(n). This condition may beexpressed as: $\begin{matrix}{\alpha_{n} = {\frac{H\left( \ell_{n,\max} \right)}{L_{ave}}.}} & {{Eq}\quad(29)}\end{matrix}$where σ² is the received noise power in a single transmission channel.The largest value of l, l_(n,max), can be identified by evaluating eachpossible value of l starting with 1 and terminating when equation (28)is no longer valid. For each value of l, the achievable SNR for the lbest transmission channels may be determined as shown by the leftargument of equation (28). This achievable SNR is then compared againstthe SNR, z_(n), required for that code rate r_(n).

Thus, for each code rate r_(n), each value of l (for l=1, 2, . . . ,l_(n,max)) is evaluated to determine whether the received SNR for eachof the l best transmission channels can achieve the associated setpointz_(n), if the total transmit power is (unevenly) distributed across alll channels. The largest value of l, l_(n,max), that satisfies thiscondition is the greatest number of transmission channels that may beselected for code rate r_(n) while achieving the required setpointz_(n).

The threshold, α_(n), associated with code rate r_(n) may then beexpressed as: $\begin{matrix}{{\beta(\ell)}{\frac{1}{\sum\limits_{i = 1}^{\ell}{\gamma(i)}^{- 1}}.}} & {{Eq}\quad(32)}\end{matrix}$The threshold α_(n) optimizes the throughput for code rate r_(n), whichrequires the setpoint z_(n). Since a common code rate is used for allselected transmission channels, the maximum achievable throughput,T_(n), can be computed as the throughput for each channel (which isr_(n)) times the number of selected channels, l_(n,max). The maximumachievable throughput T_(n) for setpoint z_(n) can then be expressed as:T _(n) =l _(n,max) r _(n),  Eq (30)where the unit for T_(n) is in information bits per modulation symbol.

The optimum throughput for the vector of setpoints can then be given by:T_(opt)=max {T_(n)}.  Eq (31)As the code rate increases, more information bits may be transmitted permodulation symbol. However, the required SNR also increases, whichrequires more transmit power for each selected transmission channel fora given noise variance σ². Since the total transmit power is limited,fewer transmission channels may be able to achieve the higher requiredSNR. Thus, the maximum achievable throughput for each code rate in thevector R may be computed, and the specific code rate that provides thehighest throughput may be deemed as the optimum code rate for thespecific channel conditions being evaluated. The optimum threshold,α_(opt), is then equal to the threshold α_(n) corresponding to thespecific code rate r_(n) that results in T_(opt).

In the above description, the optimum threshold α_(opt) is determinedbased on the channel gains for all transmission channels. If thereceived SNRs are available instead of the channel gains, then thereceived SNRs may be ranked and placed in a list γ(l) in order ofdecreasing SNRs, where 1≦l≦N_(T)N_(F), such that the first element inthe list γ(1)=max {γ(j,k)}, . . . , and the last element in the listγ(N_(T)N_(R))=min {γ(j,k)}. A sequence β(l) may then be determined as:$\begin{matrix}{\alpha_{n} = {\frac{\gamma\left( \ell_{n,\max} \right)}{\gamma_{ave}}.}} & {{Eq}\quad(34)}\end{matrix}$

For each code rate r_(n) (where 1≦n≦N_(Z)), the largest value of l,l_(n,max), is determined such that the received SNR for each of the lselected transmission channels is greater than or equal to theassociated setpoint z_(n). This condition may be expressed as:β(l)N _(T) N _(F) ≧z _(n).  Eq (33)Once the largest value of l, l_(n,max), is determined for code rater_(n), the threshold α_(n) associated with this code rate may bedetermined as: $\begin{matrix}{\alpha_{n} = {\frac{\gamma\left( \ell_{n,\max} \right)}{\gamma_{ave}}.}} & {{Eq}\quad(34)}\end{matrix}$The optimum threshold, α_(opt), and the optimum throughput, T_(opt), mayalso be determined as described above.

For the above description, the threshold is selected to optimizethroughput for the available transmission channels. The threshold mayalso be selected to optimize other performance criteria or metrics, andthis is within the scope of the invention.

FIG. 2B is a flow diagram of a process 240 to determine a threshold aused to select transmission channels for data transmission, inaccordance with an embodiment of the invention. Process 240 may be usedif the channel gains, received SNRs, or some other characteristics areavailable for the transmission channels. For clarity, process 240 isdescribed below for the case in which the channel gains are available,and the case in which the received SNRs are available is shown withinbrackets.

Initially, a vector of setpoints (Z=[z₁, z₂, . . . , z_(N) _(Z) ]) isdefined and a vector of code rates (R=[r₁, r₂, . . . , r_(N) _(Z) ])that supports the associated setpoints is determined, at step 250. Thechannel gains H(j,k) [or the received SNRs γ(j,k)] for all availabletransmission channels are retrieved and ranked from the best to theworst, at step 252. The sequence {tilde over (b)}(l) [or β(l)] ofpossible normalization factors is then determined based on the channelgains as shown in equation (27) [or based on the received SNRs as shownin equation (32)], at step 254.

Each available code rate is then evaluated via a loop. In the first stepof the loop, a (not yet evaluated) code rate r_(n) is identified forevaluation, at step 256. For the first pass through the loop, theidentified code rate can be the first code rate r₁ in the vector R. Forthe identified code rate r_(n), the largest value of l, l_(n,max), isdetermined such that the received SNR for each of the l besttransmission channels is greater than or equal to the setpoint z_(n)associated with the code rate r_(n) being evaluated, at step 258. Thiscan be performed by computing and satisfying the condition shown inequation (28) [or equation (33)]. The threshold α_(n) associated withsetpoint z_(n) is then determined based on the channel gain [or thereceived SNR] of channel l_(n,max) as shown in equation (29) [orequation (34)], at step 260. The maximum achievable throughput, T_(n),for setpoint z_(n) can also be determined as shown in equation (30), atstep 262.

A determination is then made whether or not all N_(Z) code rates havebeen evaluated, at step 264. If not, the process returns to step 256 andanother code rate is identified for evaluation. Otherwise, the optimumthroughput, T_(opt), and the optimum threshold, α_(opt), may bedetermined as shown in equation (31), at step 266. The process thenterminates.

In the above description, one threshold is determined for all availabletransmission channels in the communication system since the selectivechannel inversion is performed on all channels. In embodiments whereinthe transmission channels are segregated into a number of groups, onethreshold may be determined and used for each group. The threshold foreach group may be set based on various criteria, such as to optimize thethroughput for the transmission channels included in the group.

To determine the threshold for each group, the derivations describedabove may also be used. However, the list H_(j)(l) [or γ_(j)(l)] foreach group only includes the power gains [or received SNRs] for thetransmission channels included in the group. Also, the sequence {tildeover (b)}_(j)(l) [or β_(j)(l)] would include the possible normalizationfactors defined based on the channel gains [or received SNRs] of thetransmission channels in the group. The threshold α_(j,n) associatedwith code rate r_(n) for group j may then be expressed as:$\begin{matrix}{\alpha_{j,n} = {\frac{H_{j}\left( \ell_{n,\max} \right)}{L_{{ave},j}}\quad{or}\quad{\frac{\gamma_{j}\left( \ell_{n,\max} \right)}{\gamma_{{ave},j}}.}}} & {{Eq}\quad(35)}\end{matrix}$The optimum threshold α_(opt,j) for group j is equal to the thresholdα_(j,n) corresponding to the specific code rate r_(n) that results inthe optimal throughput T_(opt,j) for group j.

Each group of transmission channels may be associated with a respectivethreshold. Alternatively, a number of groups may share the samethreshold. This may be desirable, for example, if the same coding andmodulation scheme is to be used for a number of transmit antennas andthe available transmit power may be shared between these transmitantennas.

In the above description, the threshold is derived based on (unequal)distribution of the total available transmit power amongst the selectedtransmission channels to achieve similar received SNRs for thesechannels. In some other embodiments, the threshold may be derived basedon some other conditions and/or metrics. For example, the threshold maybe derived based on equal allocation of the total available transmitpower amongst the selected transmission channels (i.e., equal weightsfor the selected transmission channels). In this case, the threshold maybe selected to maximize the throughput achieved based on this equaltransmit power allocation. As another example, the threshold may simplybe a particular (fixed) target SNR.

Multi-Channel Communication System

FIG. 3 is a diagram of a MIMO communication system 300 capable ofimplementing various aspects and embodiments of the invention. System300 includes a first system 310 (e.g., base station 104 in FIG. 1) incommunication with a second system 350 (e.g., terminal 106). System 300may be operated to employ a combination of antenna, frequency, andtemporal diversity to increase spectral efficiency, improve performance,and enhance flexibility.

At system 310, a data source 312 provides data (i.e., information bits)to a transmit (TX) data processor 314, which (1) encodes the data inaccordance with a particular encoding scheme, (2) interleaves (i.e.,reorders) the encoded data based on a particular interleaving scheme,(3) maps the interleaved bits into modulation symbols for one or moretransmission channels selected for use for data transmission, and (4)weights the modulation symbols for each selected transmission channel.The encoding increases the reliability of the data transmission. Theinterleaving provides time diversity for the coded bits, permits thedata to be transmitted based on an average SNR for the selectedtransmission channels, combats fading, and further removes correlationbetween coded bits used to form each modulation symbol. The interleavingmay further provide frequency diversity if the coded bits aretransmitted over multiple frequency subchannels. The weightingeffectively controls the transmit power for each selected transmissionchannel to achieve a desired SNR at the receiver system. In an aspect,the coding, symbol mapping, and weighting may be performed based oncontrol signals provided by a controller 334.

A TX channel processor 320 receives and demultiplexes the weightedmodulation symbols from TX data processor 314 and provides a stream ofweighted modulation symbols for each selected transmission channel, oneweighted modulation symbol per time slot. TX channel processor 320 mayfurther precondition the weighted modulation symbols for the selectedtransmission channels if full CSI is available.

If OFDM is not employed, TX channel processor 320 provides a stream ofweighted modulation symbols for each antenna used for data transmission.And if OFDM is employed, TX channel processor 320 provides a stream ofweighted modulation symbol vectors for each antenna used for datatransmission. And if full-CSI processing is performed, TX channelprocessor 320 provides a stream of preconditioned modulation symbols orpreconditioned modulation symbol vectors for each antenna used for datatransmission. Each stream is then received and modulated by a respectivemodulator (MOD) 322 and transmitted via an associated antenna 324.

At receiver system 350, a number of receive antennas 352 receive thetransmitted signals and provide the received signals to respectivedemodulators (DEMOD) 354. Each demodulator 354 performs processingcomplementary to that performed at modulator 322. The modulation symbolsfrom all demodulators 354 are provided to a receive (RX) channel/dataprocessor 356 and processed to recover the transmitted data streams. RXchannel/data processor 356 performs processing complementary to thatperformed by TX data processor 314 and TX channel processor 320 andprovides decoded data to a data sink 360. The processing by receiversystem 350 is described in further detail below.

MIMO Transmitter Systems

FIG. 4A is a block diagram of a MIMO transmitter system 310 a, which iscapable of processing data in accordance with an embodiment of theinvention. Transmitter system 310 a is one embodiment of the transmitterportion of system 310 in FIG. 3. System 310 a includes (1) a TX dataprocessor 314 a that receives and processes information bits to provideweighted modulation symbols and (2) a TX channel processor 320 a thatdemultiplexes the modulation symbols for the selected transmissionchannels.

In the embodiment shown in FIG. 4A, TX data processor 314 a includes anencoder 412, a channel interleaver 414, a puncturer 416, a symbolmapping element 418, and a symbol weighting element 420. Encoder 412receives the aggregate information bits to be transmitted and encodesthe received bits in accordance with a particular encoding scheme toprovide coded bits. Channel interleaver 414 interleaves the coded bitsbased on a particular interleaving scheme to provide diversity.Puncturer 416 punctures (i.e., deletes) zero or more of the interleavedcoded bits to provide the desired number of coded bits. Symbol mappingelement 418 maps the unpunctured bits into modulation symbols for theselected transmission channels. And symbol weighting element 420 weighsthe modulation symbols for each selected transmission channel to provideweighted modulation symbols. The weight used for each selectedtransmission channel may be determined based on that channel's achievedSNR, as described above.

Pilot data (e.g., data of known pattern) may also be encoded andmultiplexed with the processed information bits. The processed pilotdata may be transmitted (e.g., in a time division multiplexed (TDM)manner) in a subset or all of the selected transmission channels, or ina subset or all of the available transmission channels. The pilot datamay be used at the receiver to perform channel estimation, as describedbelow.

As shown in FIG. 4A, the data encoding, interleaving, and puncturing maybe achieved based on one or more coding control signals, which identifythe specific coding, interleaving, and puncturing schemes to be used.The symbol mapping may be achieved based on a modulation control signalthat identifies the specific modulation scheme to be used. And thesymbol weighting may be achieved based on weights provided for theselected transmission channels.

In one coding and modulation scheme, the coding is achieved by using afixed base code and adjusting the puncturing to achieve the desired coderate, as supported by the SNR of the selected transmission channels. Thebase code may be a Turbo code, a convolutional code, a concatenatedcode, or some other code. The base code may also be of a particular rate(e.g., a rate ⅓ code). For this scheme, the puncturing may be performedafter the channel interleaving to achieve the desired code rate for theselected transmission channels.

Symbol mapping element 416 can be designed to group sets of unpuncturedbits to form non-binary symbols, and to map each non-binary symbol intoa point in a signal constellation corresponding to the modulation schemeselected for use for the selected transmission channels. The modulationscheme may be QPSK, M-PSK, M-QAM, or some other scheme. Each mappedsignal point corresponds to a modulation symbol.

The encoding, interleaving, puncturing, and symbol mapping attransmitter system 310 a can be performed based on numerous schemes. Onespecific scheme is described in the aforementioned U.S. patentapplication Ser. No. 09/776,075.

The number of information bits that may be transmitted for eachmodulation symbol for a particular level of performance (e.g., onepercent packet error rate or PER) is dependent on the received SNR.Thus, the coding and modulation scheme for the selected transmissionchannels may be determined based on the characteristics of the channels(e.g., the channel gains, received SNRs, or some other information). Thechannel interleaving may also be adjusted based on the coding controlsignal.

Table 1 lists various combinations of coding rate and modulation schemethat may be used for a number of received SNR ranges. The supported bitrate for each transmission channel may be achieved using any one of anumber of possible combinations of coding rate and modulation scheme.For example, one information bit per modulation symbol may be achievedusing (1) a coding rate of ½ and QPSK modulation, (2) a coding rate of ⅓and 8-PSK modulation, (3) a coding rate of ¼ and 16-QAM, or some othercombination of coding rate and modulation scheme. In Table 1, QPSK,16-QAM, and 64-QAM are used for the listed SNR ranges. Other modulationschemes such as 8-PSK, 32-QAM, 128-QAM, and so on, may also be used andare within the scope of the invention. TABLE 1 Received SNR # ofInformation Modulation # of Coded Coding Range Bits/Symbol SymbolBits/Symbol Rate 1.5-4.4 1 QPSK 2 ½ 4.4-6.4 1.5 QPSK 2 ¾  6.4-8.35 216-QAM 4 ½ 8.35-10.4 2.5 16-QAM 4 ⅝ 10.4-12.3 3 16-QAM 4 ¾  12.3-14.153.5 64-QAM 6   7/12 14.15-15.55 4 64-QAM 6 ⅔ 15.55-17.35 4.5 64-QAM 6¾ >17.35 5 64-QAM 6 ⅚

The weighted modulation symbols from TX data processor 314 a areprovided to TX channel processor 320 a, which is one embodiment of TXchannel processor 320 in FIG. 3. Within TX channel processor 320 a, ademultiplexer 424 receives and demultiplexes the weighted modulationsymbol into a number of modulation symbol streams, one stream for eachtransmission channel selected to transmit the modulation symbols. Eachmodulation symbol stream is provided to a respective modulator 322. IfOFDM is employed, the weighted modulation symbols at each time slot forall selected frequency subchannels of each transmit antenna are combinedinto a weighted modulation symbol vector. Each modulator 322 convertsthe weighted modulation symbols (for a system without OFDM) or theweighted modulation symbol vectors (for a system with OFDM) into ananalog signal, and further amplifies, filters, quadrature modulates, andupconverts the signal to generate a modulated signal suitable fortransmission over the wireless link.

FIG. 4B is a block diagram of a MIMO transmitter system 310 b, which iscapable of processing data in accordance with another embodiment of theinvention. Transmitter system 310 b is another embodiment of thetransmitter portion of system 310 in FIG. 3 and includes a TX dataprocessor 314 b and a TX channel processor 320 b.

In the embodiment shown in FIG. 4B, TX data processor 314 b includesencoder 412, channel interleaver 414, symbol mapping element 418, andsymbol weighting element 420. Encoder 412 receives and encodes theaggregate information bits in accordance with a particular encodingscheme to provide coded bits. The coding may be achieved based on aparticular code and code rate selected by controller 334, as identifiedby the coding control signals. Channel interleaver 414 interleaves thecoded bits, and symbol mapping element 418 maps the interleaved bitsinto modulation symbols for the selected transmission channels. Symbolweighting element 420 weighs the modulation symbols for each selectedtransmission channel based on a respective weight to provide weightedmodulation symbols.

In the embodiment shown in FIG. 4B, transmitter system 310 b is capableof preconditioning the weighted modulation symbols based on full CSI.Within TX channel processor 320 b, a channel MIMO processor 422demultiplexes the weighted modulation symbols into a number of (up toN_(C)) weighted modulation symbol streams, one stream for each spatialsubchannel (i.e., eigenmode) used to transmit the modulation symbols.For full-CSI processing, channel MIMO processor 422 preconditions the(up to N_(C)) weighted modulation symbols at each time slot to generateN_(T) preconditioned modulation symbols, as follows: $\begin{matrix}{\begin{bmatrix}x_{1} \\x_{2} \\\vdots \\x_{N_{T}}\end{bmatrix} = {\begin{bmatrix}{e_{11},} & {e_{12},} & \cdots & e_{1N_{C}} \\{e_{21},} & {e_{22},} & \quad & e_{2N_{C}} \\\vdots & \quad & ⋰ & \vdots \\{e_{N_{T}1},} & {e_{N_{T}1},} & \cdots & e_{N_{T}N_{C}}\end{bmatrix} \cdot \begin{bmatrix}b_{1} \\b_{2} \\\vdots \\b_{N_{C}}\end{bmatrix}}} & {{Eq}\quad(36)}\end{matrix}$where b₁, b₂, . . . b_(N) _(C) are respectively the weighted modulationsymbols for spatial

-   -   subchannels 1, 2, . . . N_(C);    -   e_(ij) are elements of an eigenvector matrix E related to the        transmission characteristics from the transmit antennas to the        receive antennas; and    -   x₁, x₂, . . . x_(N) _(T) are the preconditioned modulation        symbols, which can be expressed as:        x ₁ =b ₁ ·e ₁₁ +b ₂ ·e ₁₂ + . . . +b _(N) _(C) ·e _(1N) _(C) ,        x ₂ =b ₁ ·e ₂₁ +b ₂ ·e ₂₂ + . . . +b _(N) _(C) ·e _(2N) _(C) ,        and        x _(N) _(T) =b ₁ ·e _(N) _(T) ₁ +b ₂ ·e _(N) _(T) ₂ + . . . +b        _(N) _(C) ·e _(N) _(T) _(N) _(C) .        The eigenvector matrix E may be computed by the transmitter or        is provided to the transmitter by the receiver. The elements of        the matrix E are also taken into account in determining the        effective channel gains H(j,k).

For full-CSI processing, each preconditioned modulation symbol, x_(i),for a particular transmit antenna represents a linear combination of theweighted modulation symbols for up to N_(C) spatial subchannels. Foreach time slot, the (up to) N_(T) preconditioned modulation symbolsgenerated by channel MIMO processor 422 are demultiplexed bydemultiplexer 424 and provided to (up to) N_(T) modulators 322. Eachmodulator 322 converts the preconditioned modulation symbols (for asystem without OFDM) or the preconditioned modulation symbol vectors(for a system with OFDM) into a modulated signal suitable fortransmission over the wireless link.

FIG. 4C is a block diagram of a MIMO transmitter system 310 c, whichutilizes OFDM and is capable of processing data in accordance with yetanother embodiment of the invention. Transmitter system 310 c is anotherembodiment of the transmitter portion of system 310 in FIG. 3 andincludes a TX data processor 314 c and a TX channel processor 320 c. TXdata processor 314 c may be operated to independently code and modulateeach group of transmission channels based on a particular coding andmodulation scheme selected for the group. Each group may correspond toone transmit antenna and the transmission channels in each group maycorrespond to the frequency subchannels for the transmit antenna.

In the embodiment shown in FIG. 4C, TX data processor 314 c includes anumber of spatial subchannel data processor 410 a through 410 t, onedata processor 410 for each group of transmission channels to beindependently coded and modulated. Each data processor 410 includesencoder 412, channel interleaver 414, symbol mapping element 418, andsymbol weighting element 420. These elements of data processor 410operate to encode the information bits for a group being processed bythe data processor, interleave the coded bits, map the interleaved bitsto generated modulation symbols, and weight the modulation symbols foreach selected transmission channel within the group. As shown in FIG.4C, the coding and modulation control and the weights may bespecifically provided for each group.

The weighted modulation symbols from each data processor 410 areprovided to a respective combiner 434 within TX channel processor 320 c,which combines the weighted modulation symbols for a particular transmitantenna. If each group includes the selected frequency subchannels for aparticular transmit antenna, then combiner 434 combines the weightedmodulation symbols for the selected frequency subchannels to form amodulation symbol vector for each transmission channel, which is thenprovided to a respective modulator 322. The processing by each modulator322 to generate a modulated signal is described below.

FIG. 4D is a block diagram of a MIMO transmitter system 310 d, whichalso utilizes OFDM and is capable of processing data in accordance withyet another embodiment of the invention. In this embodiment, thetransmission channels for each frequency subchannel may be independentlyprocessed. Within a TX data processor 314 c, the information bits to betransmitted are demultiplexed by a demultiplexer 428 into a number of(up to N_(L)) frequency subchannel data streams, one stream for each ofthe frequency subchannels to be used for data transmission. Eachfrequency subchannel data stream is provided to a respective frequencysubchannel data processor 430.

Each data processor 430 processes data for a respective frequencysubchannel of the OFDM system. Each data processor 430 may beimplemented similar to TX data processor 314 a in FIG. 4A, TX dataprocessor 314 b shown in FIG. 4B, or with some other design. In oneembodiment, data processor 430 demultiplexes the frequency subchanneldata stream into a number of data substreams, one data substream foreach spatial subchannel selected for use for the frequency subchannel.Each data substream is then encoded, interleaved, symbol mapped, andweighted to generate weighted modulation symbols for the data substream.The coding and modulation for each frequency subchannel data stream oreach data substream may be adjusted based on the coding and modulationcontrol signals and the weighting may be performed based on the weights.Each data processor 430 thus provides up to N_(C) weighted modulationsymbol streams for up to N_(C) spatial subchannels selected for use forthe frequency subchannel.

For a MIMO system utilizing OFDM, the modulation symbols may betransmitted on multiple frequency subchannels and from multiple transmitantennas. Within a MIMO processor 320 d, the up to N_(C) modulationsymbol streams from each data processor 430 are provided to a respectivesubchannel spatial processor 432, which processes the receivedmodulation symbols based on the channel control and/or the availableCSI. Each spatial processor 432 may simply implement a demultiplexer(such as that shown in FIG. 4A) if full-CSI processing is not performed,or may implement a channel MIMO processor followed by a demultiplexer(such as that shown in FIG. 4B) if full-CSI processing is performed. Fora MIMO system utilizing OFDM, the full-CSI processing (i.e.,preconditioning) may be performed on each frequency subchannel.

Each subchannel spatial processor 432 demultiplexes the up to N_(C)modulation symbols for each time slot into up to N_(T) modulationsymbols for the transmit antennas selected for use for that frequencysubchannel. For each transmit antenna, a combiner 434 receives themodulation symbols for up to N_(L) frequency subchannels selected foruse for that transmit antenna, combines the symbols for each time slotinto a modulation symbol vector V, and provides the modulation symbolvector to the next processing stage (i.e., a respective modulator 322).

MIMO processor 320 d thus receives and processes the modulation symbolsto provide up to N_(T) modulation symbol vectors, V₁ through V_(Nt), onemodulation symbol vector for each transmit antenna selected for use fordata transmission. Each modulation symbol vector V covers a single timeslot, and each element of the modulation symbol vector V is associatedwith a specific frequency subchannel having a unique subcarrier on whichthe modulation symbol is conveyed.

FIG. 4D also shows an embodiment of modulator 322 for OFDM. Themodulation symbol vectors V₁ through V_(Nt) from MIMO processor 320 care provided to modulators 322 a through 322 t, respectively. In theembodiment shown in FIG. 4D, each modulator 322 includes an inverse FastFourier Transform (IFFT) 440, a cyclic prefix generator 442, and anupconverter 444.

IFFT 440 converts each received modulation symbol vector into itstime-domain representation (which is referred to as an OFDM symbol)using IFFT. IFFT 440 can be designed to perform the IFFT on any numberof frequency subchannels (e.g., 8, 16, 32, and so on). In an embodiment,for each modulation symbol vector converted to an OFDM symbol, cyclicprefix generator 442 repeats a portion of the time-domain representationof the OFDM symbol to form a “transmission symbol” for a specifictransmit antenna. The cyclic prefix insures that the transmission symbolretains its orthogonal properties in the presence of multipath delayspread, thereby improving performance against deleterious path effects.The implementation of IFFT 440 and cyclic prefix generator 442 is knownin the art and not described in detail herein.

The time-domain representations from each cyclic prefix generator 442(i.e., the transmission symbols for each antenna) are then processed(e.g., converted into an analog signal, modulated, amplified, andfiltered) by upconverter 444 to generate a modulated signal, which isthen transmitted from a respective antenna 324.

OFDM modulation is described in further detail in a paper entitled“Multicarrier Modulation for Data Transmission: An Idea Whose Time HasCome,” by John A. C. Bingham, IEEE Communications Magazine, May 1990,which is incorporated herein by reference.

FIGS. 4A through 4D show four designs of a MIMO transmitter capable ofimplementing various aspects and embodiments of the invention. Theinvention may also be practiced in an OFDM system that does not utilizeMIMO. In this case, the available transmission channels correspond tothe frequency subchannels of the OFDM system. Numerous other transmitterdesigns are also capable of implementing various inventive techniquesdescribed herein, and these designs are also within the scope of theinvention. Some of these transmitter designs are described in furtherdetail in the following patent applications, which are all assigned tothe assignee of the present application and incorporated herein byreference:

-   -   U.S. patent application Ser. No. 09/776,075, described above;    -   U.S. patent application Ser. No. 09/532,492, entitled “HIGH        EFFICIENCY, HIGH PERFORMANCE COMMUNICATIONS SYSTEM EMPLOYING        MULTI-CARRIER MODULATION,” filed Mar. 22, 2000;    -   U.S. patent application Ser. No. 09/826,481, “METHOD AND        APPARATUS FOR UTILIZING CHANNEL STATE INFORMATION IN A WIRELESS        COMMUNICATION SYSTEM,” filed Mar. 23, 2001; and    -   U.S. patent application Ser. No. 09/854,235, entitled “METHOD        AND APPARATUS FOR PROCESSING DATA IN A MULTIPLE-INPUT        MULTIPLE-OUTPUT (MIMO) COMMUNICATION SYSTEM UTILIZING CHANNEL        STATE INFORMATION,” filed May 11, 2001.        These patent applications also describe MIMO processing and CSI        processing in further detail.

In general, transmitter system 310 codes and modulates data for allselected transmission channels (or all selected transmission channelswithin each group) based a particular common coding and modulationscheme. The modulation symbols are further weighted by weights assignedto the selected transmission channels such that the desired level ofperformance is achieved at the receiver. The techniques described hereinare applicable for multiple parallel transmission channels supported byMIMO, OFDM, or any other communication scheme (e.g., a CDMA scheme)capable of supporting multiple parallel transmission channels.

FIG. 4C shows an embodiment wherein the data for each transmit antennamay be coded and modulated separately based on a coding and modulationscheme selected for that transmit antenna. Analogously, FIG. 4D shows anembodiment wherein the data for each frequency subchannel may be codedand modulated separately based on a coding and modulation schemeselected for that frequency subchannel. In general, all availabletransmission channels (e.g., all spatial subchannels of all frequencysubchannels) may be segregated into any number of groups of any type,and each group may include any number of transmission channels. Forexample, each group may include spatial subchannels, frequencysubchannels, or subchannels in both domains.

MIMO Receiver Systems

FIG. 5 is a block diagram of a MIMO receiver system 350 a capable ofreceiving data in accordance with an embodiment of the invention.Receiver system 350 a is one specific embodiment of receiver system 350in FIG. 3 and implements the successive cancellation receiver processingtechnique to receive and recover the transmitted signals. Thetransmitted signals from (up to) N_(T) transmit antennas are received byeach of N_(R) antennas 352 a through 352 r and routed to a respectivedemodulator (DEMOD) 354 (which is also referred to as a front-endprocessor).

Each demodulator 354 conditions (e.g., filters and amplifies) arespective received signal, downconverts the conditioned signal to anintermediate frequency or baseband, and digitizes the downconvertedsignal to provide samples. Each demodulator 354 may further demodulatethe samples with a received pilot to generate a stream of receivedmodulation symbols, which is provided to an RX channel/data processor356 a.

If OFDM is employed for the data transmission, each demodulator 354further performs processing complementary to that performed by modulator322 shown in FIG. 4D. In this case, each demodulator 354 includes an FFTprocessor (not shown) that generates transformed representations of thesamples and provides a stream of modulation symbol vectors. Each vectorincludes up to N_(L) modulation symbols for up to N_(L) frequencysubchannels selected for use, and one vector is provided for each timeslot. For a transmit processing scheme in which each frequencysubchannel is independently processed (e.g., as shown in FIG. 4D), themodulation symbol vector streams from the FFT processors of all N_(R)demodulators are provided to a demultiplexer (not shown in FIG. 5),which “channelizes” the modulation symbol vector stream from each FFTprocessor into up to N_(L) modulation symbol streams corresponding tothe number of frequency subchannels used for the data transmission. Thedemultiplexer then provides each of up to N_(L) modulation symbolstreams to a respective RX MIMO/data processor 356 a.

For a MIMO system not utilizing OFDM, one RX MIMO/data processor 356 amay be used to process the N_(R) modulation symbol streams from theN_(R) received antennas. And for a MIMO system utilizing OFDM, one RXMIMO/data processor 356 a may be used to process the set of N_(R)modulation symbol streams from the N_(R) received antennas for each ofup to N_(L) frequency subchannels used for data transmission.Alternatively, a single RX channel/data processor 356 a may be used toseparately process the set of modulation symbol streams associated witheach frequency subchannel.

In the embodiment shown in FIG. 5, RX channel/data processor 356 a(which is one embodiment of RX channel/data processor 356 in FIG. 3)includes a number of successive (i.e., cascaded) receiver processingstages 510, one stage for each of the transmitted data streams to berecovered by receiver system 350 a. In one transmit processing scheme,selective channel inversion is applied to all available transmissionchannels. In this case, the selected transmission channels may be usedto transmit one or more data streams, each of which may be independentlycoded with the common coding scheme. In another transmit processingscheme, selective channel inversion is applied separately to eachtransmit antenna. In this case, the selected transmission channels foreach transmit antenna may be used to transmit one or more data streams,each of which may be independently coded with the coding scheme selectedfor that transmit antenna. In general, if one data stream isindependently coded and transmitted on each spatial subchannel, then thesuccessive cancellation receiver processing technique may be used torecover the transmitted data streams. For clarity, RX channel/dataprocessor 356 a is described for an embodiment wherein one data streamis independently coded and transmitted on each spatial subchannel of agiven frequency subchannel being processed data processor 356 a.

Each receiver processing stage 510 (except for the last stage 510 n)includes a channel MIMO/data processor 520 coupled to an interferencecanceller 530, and the last stage 510 n includes only channel MIMO/dataprocessor 520 n. For the first receiver processing stage 510 a, channelMIMO/data processor 520 a receives and processes the N_(R) modulationsymbol streams from demodulators 354 a through 354 r to provide adecoded data stream for the first transmission channel (or the firsttransmitted signal). And for each of the second through last stages 510b through 510 n, channel MIMO/data processor 520 for that stage receivesand processes the N_(R) modified symbol streams from the interferencecanceller 520 in the preceding stage to derive a decoded data stream forthe transmission channel being processed by that stage. Each channelMIMO/data processor 520 further provides CSI (e.g., the received SNR)for the associated transmission channel.

For the first receiver processing stage 510 a, interference canceller530 a receives the N_(R) modulation symbol streams from all N_(R)demodulators 354. And for each of the second through second-to-laststages, interference canceller 530 receives the N_(R) modified symbolstreams from the interference canceller in the preceding stage. Eachinterference canceller 530 also receives the decoded data stream fromchannel MIMO/data processor 520 within the same stage, and performs theprocessing (e.g., coding, interleaving, modulation, channel response,and so on) to derive N_(R) remodulated symbol streams that are estimatesof the interference components of the received modulation symbol streamsdue to this decoded data stream. The remodulated symbol streams are thensubtracted from the received modulation symbol streams to derive N_(R)modified symbol streams that include all but the subtracted (i.e.,canceled) interference components. The N_(R) modified symbol streams arethen provided to the next stage.

In FIG. 5, a controller 540 is shown coupled to RX channel/dataprocessor 356 a and may be used to direct various steps in thesuccessive cancellation receiver processing performed by processor 356a.

FIG. 5 shows a receiver structure that may be used in a straightforwardmanner when each data stream is transmitted over a respective transmitantenna (i.e., one data stream corresponding to each transmittedsignal). In this case, each receiver processing stage 510 may beoperated to recover one of the transmitted signals targeted for receiversystem 350 a and provide the decoded data stream corresponding to therecovered transmitted signal.

For some other transmit processing schemes, a data stream may betransmitted over multiple transmit antennas, frequency subchannels,and/or time intervals to provide spatial, frequency, and time diversity,respectively. For these schemes, the receiver processing initiallyderives a received modulation symbol stream for the signal transmittedon each transmit antenna of each frequency subchannel. Modulationsymbols for multiple transmit antennas, frequency subchannels, and/ortime intervals may then be combined in a complementary manner as thedemultiplexing performed at the transmitter system. The stream ofcombined modulation symbols is then processed to provide thecorresponding decoded data stream.

FIG. 6A is a block diagram of an embodiment of channel MIMO/dataprocessor 520 x, which is one embodiment of channel MIMO/data processor520 in FIG. 5. In this embodiment, channel MIMO/data processor 520 xincludes a spatial/space-time processor 610, a CSI processor 612, aselector 614, a demodulation element 618, a de-interleaver 618, and adecoder 620.

Spatial/space-time processor 610 performs linear spatial processing onthe N_(R) received signals for a non-dispersive MIMO channel (i.e., withflat fading) or space-time processing on the N_(R) received signals fora dispersive MIMO channel (i.e., with frequency selective fading). Thespatial processing may be achieved using linear spatial processingtechniques such as a channel correlation matrix inversion (CCMI)technique, a minimum mean square error (MMSE) technique, and others.These techniques may be used to null out the undesired signals or tomaximize the received SNR of each of the constituent signals in thepresence of noise and interference from the other signals. Thespace-time processing may be achieved using linear space-time processingtechniques such as a MMSE linear equalizer (MMSE-LE), a decisionfeedback equalizer (DFE), a maximum-likelihood sequence estimator(MLSE), and others. The CCMI, MMSE, MMSE-LE, and DFE techniques aredescribed in further detail in the aforementioned U.S. patentapplication Ser. No. 09/854,235. The DFE and MLSE techniques are alsodescribed in further detail by S. L. Ariyavistakul et al. in a paperentitled “Optimum Space-Time Processors with Dispersive Interference:Unified Analysis and Required Filter Span,” IEEE Trans. onCommunication, Vol. 7, No. 7, July 1999, and incorporated herein byreference.

CSI processor 612 determines the CSI for each of the transmissionchannels used for data transmission. For example, CSI processor 612 mayestimate a noise covariance matrix based on the received pilot signalsand then compute the SNR of the k-th transmission channel used for thedata stream to be decoded. The SNR may be estimated similar toconventional pilot assisted single and multi-carrier systems, as isknown in the art. The SNR for all of the transmission channels used fordata transmission may comprise the CSI that is reported back to thetransmitter system. CSI processor 612 may further provide to selector614 a control signal that identifies the particular data stream to berecovered by this receiver processing stage.

Selector 614 receives a number of symbol streams from spatial/space-timeprocessor 610 and extracts the symbol stream corresponding to the datastream to be decoded, as indicated by the control signal from CSIprocessor 612. The extracted stream of modulation symbols is thenprovided to a demodulation element 614.

For the embodiment shown in FIG. 6A in which the data stream for eachtransmission channel is independently coded and modulated based on thecommon coding and modulation scheme, the recovered modulation symbolsfor the selected transmission channel are demodulated in accordance witha demodulation scheme (e.g., M-PSK, M-QAM) that is complementary to thecommon modulation scheme used for the transmission channel. Thedemodulated data from demodulation element 616 is then de-interleaved bya de-interleaver 618 in a complementary manner to that performed bychannel interleaver 614, and the de-interleaved data is further decodedby a decoder 620 in a complementary manner to that performed by encoder612. For example, a Turbo decoder or a Viterbi decoder may be used fordecoder 620 if Turbo or convolutional coding, respectively, is performedat the transmitter system. The decoded data stream from decoder 620represents an estimate of the transmitted data stream being recovered.

FIG. 6B is a block diagram of an interference canceller 530 x, which isone embodiment of interference canceller 530 in FIG. 5. Withininterference canceller 530 x, the decoded data stream from the channelMIMO/data processor 520 within the same stage is re-encoded,interleaved, and re-modulated by a channel data processor 628 to provideremodulated symbols, which are estimates of the modulation symbols atthe transmitter system prior to the MIMO processing and channeldistortion. Channel data processor 628 performs the same processing(e.g., encoding, interleaving, and modulation) as that performed at thetransmitter system for the data stream. The remodulated symbols are thenprovided to a channel simulator 630, which processes the symbols withthe estimated channel response to provide an estimate, î^(k), of theinterference due the decoded data stream. The channel response estimatemay be derived based on the pilot and/or data transmitted by thetransmitter system and in accordance with the techniques described inthe aforementioned U.S. patent application Ser. No. 09/854,235.

The N_(R) elements in the interference vector î^(k) correspond to thecomponent of the received signal at each of the N_(R) receive antennasdue to symbol stream transmitted on the k-th transmit antenna. Eachelement of the vector represents an estimated component due to thedecoded data stream in the corresponding received modulation symbolstream. These components are interference to the remaining (not yetdetected) transmitted signals in the N_(R) received modulation symbolstreams (i.e., the vector r^(k)), and are subtracted (i.e., canceled)from the received signal vector r^(k) by a summer 632 to provide amodified vector r^(k+1) having the components from the decoded datastream removed. The modified vector r^(k+1) is provided as the inputvector to the next receiver processing stage, as shown in FIG. 5.

Various aspects of the successive cancellation receiver processing aredescribed in further detail in the aforementioned U.S. patentapplication Ser. No. 09/854,235.

FIG. 7 is a block diagram of a MIMO receiver system 350 b capable ofreceiving data in accordance with another embodiment of the invention.The transmitted signals from (up to) N_(T) transmit antennas arereceived by each of N_(R) antennas 352 a through 352 r and routed to arespective demodulator 354. Each demodulator 354 conditions, processes,and digitizes a respective received signal to provide samples, which areprovided to a RX MIMO/data processor 356 b.

Within RX MIMO/data processor 356 b, the samples for each receiveantenna are provided to a respective FFT processor 710, which generatestransformed representations of the received samples and provides arespective stream of modulation symbol vectors. The streams ofmodulation symbol vector from FFT processors 710 a through 710 r arethen provided to a processor 720. Processor 720 channelizes the streamof modulation symbol vectors from each FFT processor 710 into a numberof up to N_(L) subchannel symbol streams. Processor 720 may furtherperform spatial processing or space-time processing on the subchannelsymbol streams to provide post-processed modulation symbols.

For each data stream transmitted over multiple frequency subchannelsand/or multiple spatial subchannels, processor 720 further combines themodulation symbols for all frequency and spatial subchannels used fortransmitting the data stream into one post-processed modulation symbolstream, which is then provided to a data stream processor 730. Each datastream processor 730 performs demodulation, de-interleaving, anddecoding complementary to that performed on the data stream at thetransmitter unit and provides a respective decoded data stream.

Receiver systems that employ the successive cancellation receiverprocessing technique and those that do not employ the successivecancellation receiver processing technique may be used to receive,process, and recover the transmitted data streams. Some receiver systemscapable of processing signals received over multiple transmissionchannels are described in the aforementioned U.S. patent applicationSer. Nos. 09/776,075 and 09/826,481, and U.S. patent application Ser.No. 09/532,492, entitled “HIGH EFFICIENCY, HIGH PERFORMANCECOMMUNICATIONS SYSTEM EMPLOYING MULTI-CARRIER MODULATION,” filed Mar.30, 2000, assigned to the assignee of the present invention andincorporated herein by reference.

Obtaining CSI for the Transmitter System

For simplicity, various aspects and embodiments of the invention havebeen described wherein the CSI comprises SNR. In general, the CSI maycomprise any type of information that is indicative of thecharacteristics of the communication link. Various types of informationmay be provided as CSI, some examples of which are described below.

In one embodiment, the CSI comprises SNR, which is derived as the ratioof the signal power over the noise plus interference power. The SNR istypically estimated and provided for each transmission channel used fordata transmission (e.g., each transmit data stream), although anaggregate SNR may also be provided for a number of transmissionchannels. The SNR estimate may be quantized to a value having aparticular number of bits. In one embodiment, the SNR estimate is mappedto an SNR index, e.g., using a look-up table.

In another embodiment, the CSI comprises power control information foreach spatial subchannel of each frequency subchannel. The power controlinformation may include a single bit for each transmission channel toindicate a request for either more power or less power, or it mayinclude multiple bits to indicate the magnitude of the change of powerlevel requested. In this embodiment, the transmitter system may make useof the power control information fed back from the receiver systems todetermine which transmission channels to select, and what power to usefor each transmission channel.

In yet another embodiment, the CSI comprises signal power andinterference plus noise power. These two components may be separatelyderived and provided for each transmission channel used for datatransmission.

In yet another embodiment, the CSI comprises signal power, interferencepower, and noise power. These three components may be derived andprovided for each transmission channel used for data transmission.

In yet another embodiment, the CSI comprises signal-to-noise ratio plusa list of interference powers for each observable interference term.This information may be derived and provided for each transmissionchannel used for data transmission.

In yet another embodiment, the CSI comprises signal components in amatrix form (e.g., N_(T)×N_(R) complex entries for all transmit-receiveantenna pairs) and the noise plus interference components in matrix form(e.g., N_(T)×N_(R) complex entries). The transmitter system may thenproperly combine the signal components and the noise plus interferencecomponents for the appropriate transmit-receive antenna pairs to derivethe quality for each transmission channel used for data transmission(e.g., the post-processed SNR for each transmitted data stream, asreceived at the receiver systems).

In yet another embodiment, the CSI comprises a data rate indicator foreach transmit data stream. The quality of a transmission channel to beused for data transmission may be determined initially (e.g., based onthe SNR estimated for the transmission channel) and a data ratecorresponding to the determined channel quality may then be identified(e.g., based on a look-up table). The identified data rate is indicativeof the maximum data rate that may be transmitted on the transmissionchannel for the required level of performance. The data rate is thenmapped to and represented by a data rate indicator (DRI), which can beefficiently coded. For example, if (up to) seven possible data rates aresupported by the transmitter system for each transmit antenna, then a3-bit value may be used to represent the DRI where, e.g., a zero mayindicate a data rate of zero (i.e., don't use the transmit antenna) and1 through 7 may be used to indicate seven different data rates. In atypical implementation, the quality measurements (e.g., SNR estimates)are mapped directly to the DRI based on, e.g., a look-up table.

In yet another embodiment, the CSI comprises an indication of theparticular processing scheme to be used at the transmitter system foreach transmit data stream. In this embodiment, the indicator mayidentify the particular coding scheme and the particular modulationscheme to be used for the transmit data stream such that the desiredlevel of performance is achieved.

In yet another embodiment, the CSI comprises a differential indicatorfor a particular measure of quality for a transmission channel.Initially, the SNR or DRI or some other quality measurement for thetransmission channel is determined and reported as a referencemeasurement value. Thereafter, monitoring of the quality of thetransmission channel continues, and the difference between the lastreported measurement and the current measurement is determined. Thedifference may then be quantized to one or more bits, and the quantizeddifference is mapped to and represented by the differential indicator,which is then reported. The differential indicator may indicate toincrease or decrease the last reported measurement by a particular stepsize (or to maintain the last reported measurement). For example, thedifferential indicator may indicate that (1) the observed SNR for aparticular transmission channel has increased or decreased by aparticular step size, or (2) the data rate should be adjusted by aparticular amount, or some other change. The reference measurement maybe transmitted periodically to ensure that errors in the differentialindicators and/or erroneous reception of these indicators do notaccumulate.

In yet another embodiment, the CSI comprises the channel gain for eachavailable transmission channel, as estimated at the receiver systembased on signals transmitted by the transmitter system.

Other forms of CSI may also be used and are within the scope of theinvention. In general, the CSI includes sufficient information inwhatever form that may be used to (1) select a set of transmissionchannels that will result in optimum or near optimum throughput, (2)determine a weighting factor for each selected transmission channel thatresults in equal or near equal received SNRs, and (3) infer an optimumor near optimum code rate for the selected transmission channels.

The CSI may be derived based on the signals transmitted from thetransmitter system and received at the receiver systems. In anembodiment, the CSI is derived based on a pilot reference included inthe transmitted signals. Alternatively or additionally, the CSI may bederived based on the data included in the transmitted signals. Althoughdata may be transmitted on only the selected transmission channels,pilot data may be transmitted on unselected transmission channels toallow the receiver systems to estimate the channel characteristics.

In yet another embodiment, the CSI comprises one or more signalstransmitted from the receiver systems to the transmitter system. In somesystems, a degree of correlation may exist between the uplink anddownlink (e.g. time division duplexed (TDD) systems where the uplink anddownlink share the same frequency band in a time division multiplexedmanner). In these systems, the quality of the uplink may be estimated(to a requisite degree of accuracy) based on the quality of thedownlink, and vice versa, which may be estimated based on signals (e.g.,pilot signals) transmitted from the receiver systems. The pilot signalswould then represent a means for which the transmitter system couldestimate the CSI as observed at the receiver systems. For this type ofCSI, no reporting of channel characteristics is necessary.

The signal quality may be estimated at the transmitter system based onvarious techniques. Some of these techniques are described in thefollowing patents, which are assigned to the assignee of the presentapplication and incorporated herein by reference:

-   -   U.S. Pat. No. 5,799,005, entitled “SYSTEM AND METHOD FOR        DETERMINING RECEIVED PILOT POWER AND PATH LOSS IN A CDMA        COMMUNICATION SYSTEM,” issued Aug. 25, 1998,    -   U.S. Pat. No. 5,903,554, entitled “METHOD AND APPARATUS FOR        MEASURING LINK QUALITY IN A SPREAD SPECTRUM COMMUNICATION        SYSTEM,” issued May 11, 1999,    -   U.S. Pat. Nos. 5,056,109, and 5,265,119, both entitled “METHOD        AND APPARATUS FOR CONTROLLING TRANSMISSION POWER IN A CDMA        CELLULAR MOBILE TELEPHONE SYSTEM,” respectively issued Oct. 8,        1991 and Nov. 23, 1993, and    -   U.S. Pat. No. 6,097,972, entitled “METHOD AND APPARATUS FOR        PROCESSING POWER CONTROL SIGNALS IN CDMA MOBILE TELEPHONE        SYSTEM,” issued Aug. 1, 2000.        Methods for estimating a single transmission channel based on a        pilot signal or a data transmission may also be found in a        number of papers available in the art. One such channel        estimation method is described by F. Ling in a paper entitled        “Optimal Reception, Performance Bound, and Cutoff-Rate Analysis        of References-Assisted Coherent CDMA Communications with        Applications,” IEEE Transaction On Communication, October 1999.

Various types of information for CSI and various CSI reportingmechanisms are also described in U.S. patent application Ser. No.08/963,386, entitled “METHOD AND APPARATUS FOR HIGH RATE PACKET DATATRANSMISSION,” filed Nov. 3, 1997, assigned to the assignee of thepresent application, and in “TIE/EIA/IS-856 cdma2000 High Rate PacketData Air Interface Specification”, both of which are incorporated hereinby reference.

The CSI may be reported back to the transmitter using various CSItransmission schemes. For example, the CSI may be sent in full,differentially, or a combination thereof. In one embodiment, CSI isreported periodically, and differential updates are sent based on theprior transmitted CSI. In another embodiment, the CSI is sent only whenthere is a change (e.g., if the change exceeds a particular threshold),which may lower the effective rate of the feedback channel. As anexample, the SNRs may be sent back (e.g., differentially) only when theychange. For an OFDM system (with or without MIMO), correlation in thefrequency domain may be exploited to permit reduction in the amount ofCSI to be fed back. As an example for an OFDM system, if the SNRcorresponding to a particular spatial subchannel for M frequencysubchannels is the same, the SNR and the first and last frequencysubchannels for which this condition is true may be reported. Othercompression and feedback channel error recovery techniques to reduce theamount of data to be fed back for CSI may also be used and are withinthe scope of the invention.

Referring back to FIG. 3, the CSI (e.g., the received SNR) determined byRX channel/data processor 356 is provided to a TX data processor 362,which processes the CSI and provides processed data to one or moremodulators 354. Modulators 354 further condition the processed data andtransmit the CSI back to transmitter system 310 via a reverse channel.

At system 310, the transmitted feedback signal is received by antennas324, demodulated by demodulators 322, and provided to a RX dataprocessor 332. RX data processor 332 performs processing complementaryto that performed by TX data processor 362 and recovers the reportedCSI, which is then provided to controller 334.

Controller 334 uses the reported CSI to perform a number of functionsincluding (1) selecting the set of N_(S) best available transmissionchannels for data transmission, (2) determining the coding andmodulation scheme to be used for data transmission on the selectedtransmission channels, and (3) determining the weights to be used forthe selected transmission channels. Controller 334 may select thetransmission channels to achieve high throughput or based on some otherperformance criteria or metrics, and may further determine the thresholdused to select the transmission channels, as described above.

The characteristics (e.g., channel gains or received SNRs) of thetransmission channels available for data transmission may be determinedbased on various techniques as described above and provided to thetransmitter system. The transmitter system may then use the informationto select the set of N_(S) best transmission channels, properly code andmodulate the data, and further weight the modulation symbols.

The techniques described herein may be used for data transmission on thedownlink from a base station to one or more terminals, and may also beused for data transmission on the uplink from each of one or moreterminals to a base station. For the downlink, transmitter system 310 inFIGS. 3 and 4A through 4D may represent part of a base station andreceiver system 350 in FIGS. 3, 5, and 6 may represent part of aterminal. And for the uplink, transmitter system 310 in FIGS. 3 and 4Athrough 4D may represent part of a terminal and receiver system 350 inFIGS. 3, 5, and 6 may represent part of a base station.

The elements of the transmitter and receiver systems may be implementedwith one or more digital signal processors (DSP), application specificintegrated circuits (ASIC), processors, microprocessors, controllers,microcontrollers, field programmable gate arrays (FPGA), programmablelogic devices, other electronic units, or any combination thereof. Someof the functions and processing described herein may also be implementedwith software executed on a processor. Certain aspects of the inventionmay also be implemented with a combination of software and hardware. Forexample, computations to determine the threshold, α, and to selecttransmission channels may be performed based on program codes executedon a processor (controller 334 in FIG. 3).

Headings are included herein for reference and to aid in the locatingcertain sections. These heading are not intended to limit the scope ofthe concepts described therein under, and these concepts may haveapplicability in other sections throughout the entire specification.

The previous description of the disclosed embodiments is provided toenable any person skilled in the art to make or use the presentinvention. Various modifications to these embodiments will be readilyapparent to those skilled in the art, and the generic principles definedherein may be applied to other embodiments without departing from thespirit or scope of the invention. Thus, the present invention is notintended to be limited to the embodiments shown herein but is to beaccorded the widest scope consistent with the principles and novelfeatures disclosed herein.

1. A method for processing data for transmission over multipletransmission channels in a multi-channel communication system,comprising: determining characteristics of a plurality of transmissionchannels available for data transmission; segregating the plurality oftransmission channels into one or more groups of transmission channels;and for each group of transmission channels, selecting one or moreavailable transmission channels in the group based on the determinedcharacteristics and a threshold, and coding and modulating data for allselected transmission channels in the group based on a particular codingand modulation scheme to provide modulation symbols.
 2. A method fortransmitting data over multiple transmission channels in a multi-channelcommunication system, comprising: coding and modulating data fortransmission over transmission channels in a group of transmissionchannels of a plurality of groups of transmission channels to providemodulation symbols; weighting modulation symbols for each transmissionchannel in each group based on a respective weight indicative of atransmit power level for the transmission channel and derived based inpart on the characteristics of the transmission channel; andtransmitting the weighted modulation symbols on the selectedtransmission channels.
 3. The method of claim 2, wherein themulti-channel communication system is a multiple-input multiple-output(MIMO) that utilizes orthogonal frequency division modulation (OFDM). 4.The method of claim 2, wherein each group corresponds to a respectivetransmit antenna.
 5. The method of claim 2, wherein the plurality oftransmission channels in each group correspond to a plurality offrequency subchannels for a corresponding transmit antenna.
 6. Themethod of claim 2, wherein coding and modulating comprises coding andmodulating the data for the selected transmission channels in each groupis coded based on a common coding scheme.
 7. The method of claim 6,wherein the common coding scheme is selected from among a plurality ofpossible coding schemes.
 8. The method of claim 2, further comprisingselecting one or more of the available transmission channels in eachgroup for use for data transmission.
 9. The method of claim 8, whereinselecting comprises selecting one or more of the available transmissionchannels in each group for use for data transmission based on thecharacteristics of the transmission channels and a threshold.
 10. Themethod of claim 9, wherein each group is associated with a respectivethreshold.
 11. The method of claim 2, fiber comprising deriving theweights for the transmission channels in each group to distribute totaltransmit power available for the group among all transmission channelsin the group to achieve similar received signal quality.
 12. The methodof claim 11, wherein the received signal quality is a function of asignal-to-noise-plus-interference ratio (SNR).
 13. The method of claim2, wherein the characteristics for the available transmission channelsare channel gains.
 14. The method of claim 2, wherein thecharacteristics for the available transmission channels are receivedsignal-to-noise-plus-interference ratios (SNRs).
 15. The method of claim2, further comprising deriving the weights for the transmission channelsin each group based on a total transmit power available for the group inwhich the transmission channel belongs.
 16. The method of claim 2,further comprising deriving the weight for each selected transmissionchannel based on a normalization factor, which is determined based onthe characteristics of the selected transmission channels.
 17. Themethod of claim 2, further comprising determining the characteristics ofthe transmission channel based upon channel state information receivedfrom an intended receiver of the weighted modulation symbols.
 18. Themethod of claim 17, wherein the channel state information comprisesinformation indicative of a signal-to-noise-plus-interference ratio(SNR).
 19. The method of claim 17, wherein the multiple transmissionchannels each comprise a propagation path between each transmit-receiveantenna pair for a plurality of frequency subchannels and wherein thechannel state information comprises a characterization each transmissionchannel of the multiple transmission channels.
 20. A transmitter for usein a multi-channel communication system, comprising: a transmit dataprocessor configured to code and modulate data for transmission overtransmission channels in a group of transmission channels of a pluralityof groups of transmission channels to provide modulation symbols and toweight the modulation symbols for each transmission channel in eachgroup based on a respective weight indicative of a transmit power levelfor the transmission channel and derived based in part on thecharacteristics of the transmission channel; and a memory coupled withthe processor.
 21. The transmitter of claim 20, wherein themulti-channel communication system is a multiple-input multiple-output(MIMO) that utilizes orthogonal frequency division modulation (OFDM).22. The transmitter of claim 20, wherein each group corresponds to arespective transmit antenna of the wireless communication device. 23.The transmitter of claim 20, wherein the plurality of transmissionchannels in each group correspond to a plurality of frequencysubchannels for a corresponding transmit antenna.
 24. The transmitter ofclaim 20, wherein the transmit data process is configured to code andmodulate the data for the selected transmission channels in each groupbased on a common coding scheme.
 25. The transmitter of claim 20,further comprising a controller coupled with the processor andconfigured to select one or more of the available transmission channelsin each group for use for data transmission.
 26. The transmitter ofclaim 20, further comprising a controller coupled with the processor andconfigured to derive the weights for the transmission channels in eachgroup to distribute total transmit power available for the group amongall transmission channels in the group to achieve similar receivedsignal quality.
 27. The transmitter of claim 26, wherein the receivedsignal quality is a function of a signal-to-noise-plus-interferenceratio (SNR).
 28. The transmitter of claim 20, wherein thecharacteristics for the available transmission channels are channelgains.
 29. The transmitter of claim 20, wherein the characteristics forthe available transmission channels are receivedsignal-to-noise-plus-interference ratios (SNRs).
 30. The transmitter ofclaim 20, further comprising a controller coupled with the processor andconfigured to derive the weights for the transmission channels in eachgroup based on a total transmit power available for the group in whichthe transmission channel belongs.
 31. The transmitter of claim 20,further comprising a controller coupled with the processor andconfigured to derive the weight for each selected transmission channelbased on a normalization factor, which is determined based on thecharacteristics of the selected transmission channels.
 32. Thetransmitter of claim 20, further comprising a controller coupled withthe processor and configured to determine the characteristics of thetransmission channel based upon channel state information received froman intended receiver of the weighted modulation symbols.
 33. Thetransmitter of claim 20, wherein the channel state information comprisesinformation indicative of a signal-to-noise-plus-interference ratio(SNR).
 34. The transmitter of claim 20, wherein the multipletransmission channels each comprise a propagation path between eachtransmit-receive antenna pair for a plurality of frequency subchannelsand wherein the channel state information comprises a characterizationeach transmission channel of the multiple transmission channels.
 35. Thetransmitter of claim 20, wherein the transmitter comprises a transmitterof an access point.
 36. The transmitter of claim 20, wherein thetransmitter comprises a transmitter of an access terminal.
 37. Anapparatus for processing data for transmission over multipletransmission channels in a multi-channel communication system,comprising: means for coding and modulating data for transmission overtransmission channels in a group of transmission channels of a pluralityof groups of transmission channels to provide modulation symbols; meansfor weighting modulation symbols for each transmission channel in eachgroup based on a respective weight indicative of a transmit power levelfor the transfusion channel and derived based in part on thecharacteristics of the transmission channel; and a transmitterconfigured to transmit the weighted modulation symbols on the selectedtransmission channels.
 38. The apparatus of claim 37, wherein themulti-channel communication system is a multiple-input multiple-output(MIMO) that utilizes orthogonal frequency division modulation (OFDM).39. The apparatus of claim 37, wherein each group corresponds to arespective transmit antenna.
 40. The apparatus of claim 37, wherein theplurality of transmission channels in each group correspond to aplurality of frequency subchannels for a corresponding transmit antenna.41. The apparatus of claim 37, wherein the means for coding andmodulating comprises means for coding and modulating the data for theselected transmission channels in each group is coded based on a commoncoding scheme.
 42. The apparatus of claim 37, further comprising meansfor selecting one or more of the available transmission channels in eachgroup for use for data transmission.
 43. The apparatus of claim 37,further comprising means for deriving the weights for the transmissionchannels in each group to distribute total transmit power available forthe group among all transmission channels in the group to achievesimilar received signal quality.
 44. The apparatus of claim 37, whereinthe characteristics for the available transmission channels are channelgains.
 45. The apparatus of claim 37, wherein the characteristics forthe available transmission channels are receivedsignal-to-noise-plus-interference ratios (SNRs).
 46. The apparatus ofclaim 37, further comprising means for deriving the weights for thetransmission channels in each group based on a total transmit poweravailable for the group in which the transmission channel belongs. 47.The apparatus of claim 37, flanker comprising means for deriving theweight for each selected transmission channel based on a normalizationfactor, which is determined based on the characteristics of the selectedtransmission channels.
 48. The apparatus of claim 37, further comprisingmeans for determining the characteristics of the transmission channelbased upon channel state information received from an intended receiverof the weighted modulation symbols.
 49. A processor readable mediacomprising instructions thereon that may be utilized by a processor forprocessing data for transmission over multiple transmission channels ina multi-channel communication system, the instructions comprising:instructions for coding and modulating data for transmission overtransmission channels in a group of transmission channels of a pluralityof groups of transmission channels to provide modulation symbols; andinstructions for weighting modulation symbols for each transmissionchannel in each group based on a respective weight indicative of atransmit power level for the transmission channel and derived based inpart on the characteristics of the transmission channel.